This invention relates to a transmitting method and apparatus and, more particularly, to a transmitting method and apparatus in which signals obtained by applying an inverse Fourier transform to a plurality of digital subcarrier signals having frequencies that differ from one another are transmitted after being subjected at least to a digital-to-analog conversion.
Baseband modulation processing has become complicated in recent years as in the case of CDMA (Code Division Multiple Access) and OFDM (Orthogonal Frequency Division Multiplexing), and numerical processing often is executed using integrated circuits such as an FPGA (Field Programmable Gate Array) or LSI (Large-Scale Integration). FIG. 9 illustrates an example of the structure of a modulating apparatus for performing OFDM. This is a typical example of the prior art.
A digital signal processor 1 executes digital signal processing in conformity with a reference clock signal CLK that is output from an oscillator 2. More specifically, a baseband signal processor 1a executes signal processing (baseband signal processing) that is not essential for OFDM, e.g., processing such as appending of an error correction/detection code to a signal to be transmitted, interleaving, multivalued modulation and code spreading. A serial/parallel converter (S/P converter) 1b converts the result of processing (transmit data) by the baseband signal processor 1a to parallel data, and an IFFT (inverse fast-Fourier transform) unit 1c executes IFFT processing using N-number of items of parallel data D0 to DN−1 as N-number of subcarriers, thereby effecting a conversion to discrete time signals. Depending upon the scheme, there are also arrangements in which interleaving, multivalued modulation and code spreading are executed after the S/P conversion, unlike the above scheme. Either arrangement may be used in the present invention.
IFFT signal processing is easy to understand if it is described based upon the concept of subcarriers in the frequency domain. N-number of input signals enter the IFFT unit 1c in parallel as N-number of subcarrier components. These signals give the size and argument of the N-number of subcarriers. With a frequency that is 1/N of the FFT sampling frequency fN serving as a reference, each subcarrier is a complex sine wave that is a whole-number multiple of this reference frequency. Here N represents the size of the FFT (fast-Fourier transform) and fN the FFT sampling frequency.
The frequency of an n-th subcarrier is given by the following equation:fn=n/NfN, (n=integer of −N/2˜N/2−1)OFDM is an operation through which an output signal is obtained by adding all complex sinusoidal signals generated by these subcarriers.
Consider a more practical OFDM scheme. A number N′ of subcarriers used as shown in FIG. 10 (0 is not input for these subcarriers) is smaller than the FFT size N. Here it is so arranged that the subcarrier f0 corresponding to a DC component and subcarriers in the vicinity of f−N/2, f+N/2 are not used. Adopting this arrangement improves reliability or facilitates RF transmit-signal processing. (For example, see The Institute of Electronics, Information and Communication Engineers, 1998 Software Convention B-5-178 “Review of Higher-Harmonic Suppression Filters in OFDM Modulator”, NTT Wireless System Research Laboratory, and The Institute of Electronics, Information and Communication Engineers, 1998 Software Convention B-5-241 “Review of DC Offset in OFDM Modem”, NTT Wireless System Research Laboratory.)
FIG. 11 is a frequency spectrum in a case where subcarriers used as shown in FIG. 10 have been selected.
The signal that has undergone IFFT processing is input to a GI (Guard Interval) add-on unit 1d. The latter executes processing to add a guard interval onto the IFFT output signal in order to eliminate inter-symbol interference. The processing above is executed by an FPGA or LSI circuit and the output thereof is the result of a numerical operation. A digital analog converter (DAC) 1e is used to convert the result of this numerical operation to an actual electric signal in the form of voltage or current.
The analog baseband signal obtained by the above-described processing contains many higher harmonic components in view of the nature of IFFT or DAC processing. This signal is band-limited by a low-pass filter (LPF) 3 to extract an analog baseband signal of a desired band and input this baseband signal to a mixer (MIX) 4 such as a quadrature modulator. The mixer 4 mixes a high-frequency signal, which is generated by an oscillator 5, with the baseband signal and converts this baseband signal to a high-frequency signal. An image component produced by mixing and unwanted waves such as spurious waves are removed by a bandpass filter (BPF) 6, after which the resultant signal is transmitted from an antenna via a high-frequency amplifier (not shown) or the like as a modulated signal.
The main reason why the subcarrier f0 that corresponds to the DC component is not used (the reason why f0=DC is made zero) in the IFFT processing of FIG. 9 is to avoid a situation in which local leakage produced by the frequency conversion using the analog mixer interferes with the subcarrier f0. For example, if sine waves having the frequency fA of the carrier and the frequency fB of the baseband signal are mixed, signals representing fA+fB and fA−fB are generated and, in theory, a signal of frequency fA will not appear. In the actual circuitry, however, a leakage component of fA referred to as local leakage appears. If this leakage component were added to the subcarrier f0, the information possessed by this subcarrier would tend to be erroneous. Empirically, this local leakage is a component as low as 10 to 40 dB with respect to the carrier but it is not zero. The subcarrier f0 is not used for this reason.
Further, the reason why subcarriers in the vicinity of f−N/2, f+N/2 (e.g., a frequency domain having a width greater than at least the frequency spacing 2fN/N of the subcarriers) are not used is that if they were used, the filter 3 would be required to have a steep characteristic. That is, before the signal frequency in the band of the baseband is up-converted, the band signal of the baseband is passed using the filter 3. However, if the subcarriers in the vicinity of f−N/2, f+N/2 were to be used, the frequency components would be rendered continuous and it would become difficult to separate the baseband band signal. This would necessitate a filter having an steep characteristic.
The OFDM transmitting apparatus executes the OFDM transmit-signal processing described above. In most of these apparatus the mixing by the mixer is performed in two stages, namely in the intermediate frequency (IF) band and high-frequency (RF) band. In the present invention, however, it is not necessary to emphasize this distinction and this aspect is omitted. Further, though it is necessary to execute processing such as power amplification before a signal is transmitted from the antenna, this is not a requisite component of the present invention and is omitted. Furthermore, the order of the mixer, BPF and amplifier, variations relating to the number thereof and isolators essential between these stages are similarly omitted.
The oscillator 5 and mixer 4 of FIG. 9 have an analog circuit structure. As a consequence, cost rises in pursuit of a reduction in unwanted signal components in the mixer, broadening of the band and greater stability of the oscillator. Further, the analog oscillator and mixer are substantially fixed in terms of frequency setting. This becomes a barrier when considering software wireless techniques that require flexible changes in circuitry. For this reason, there is prior art in which analog circuitry is eliminated from the arrangement of FIG. 9 (see the specifications of JP 8-23359A, JP 8-149170A and JP 52-113668A). This prior art applies special processing to the above-described digital baseband signal, extracts a prescribed frequency component by a band-pass filter BPF and uses this component as a modulated signal whose frequency has been up-converted. This prior art is premised upon the fact that a signal sampled at a sampling frequency fs and then output through a DAC is obtained as an ideal sampling signal for which pulse width τ=0 holds, as shown in FIG. 12. If the sampling frequency fs (=1/Ts) is selected to be the Nyquist frequency (fs>2·fb, where fb represents the baseband frequency) in such an ideal case, then a repetitive higher-harmonic component is not attenuated and can be made to appear every whole-number multiple of this frequency, thereby making it possible to extract a desired frequency component from the higher-harmonic component.
In general, digital signal processing performs a numerical operation using an FPGA or LSI circuit and the signal must be realized as an electric signal using the DAC 5. The DAC output signal is such that it is difficult to reduce pulse width as the band of communication broadens, and a zero-order hold state results, as shown in FIG. 14, in which the value at a certain clock timing is maintained until the next clock timing. As a result, the spectrum of the zero-order hold signal differs from that of the ideal sampling signal, as illustrated in FIG. 15, sinx/x-shaped amplitude attenuation is sustained along the frequency axis and the signal component becomes zero at a specific frequency. This phenomenon is referred to as “zero-cross”. With a signal of the zero-order hold state where τ=Ts holds, zero-cross is produced every whole-number multiple of fs (=1/τ). As a result, the shape of the higher harmonic declines markedly in the vicinity of whole-number multiples of fs and a higher harmonic 100 to be extracted undergoes a great amount of attenuation in the vicinity of the center of the band and is difficult to utilize as a modulated signal. In other words, a higher-harmonic component of any degree can no longer be extracted. FIG. 16 is a diagram useful in describing prior art adapted so that it can be compared with the present invention, described later. In FIG. 16, (a) shows the spectrum of the baseband signal that is input to the DA converter 1e, (b) the spectrum of the baseband signal that is output from the DA converter 1e when sampling has been performed at an impulse sequence in the DA converter, (c) the spectrum of the baseband signal, which has sustained sinx/x-shaped amplitude attenuation, output from the DA converter 1e at the time of sample-and-hold, and (d) the spectrum of the baseband signal that prevails when the frequency of a receive signal 101 has been down-converted in a receiving apparatus. The signal declines markedly at the center of the band and the probability that information cannot be decoded correctly increases.
The examples of the prior art described in the specifications of JP 8-23359A and JP 8-149170A take this decline in amplitude based upon sinx/x into account but give rise to the following problem: The art described in JP 8-23359A is such that in view of the occurrence of sinx/x-shaped amplitude attenuation in which the higher harmonic obtained by a sampling operation whose duty ratio is 50% experiences zero-cross occurs at odd-numbered multiples of the fundamental frequency, the higher harmonic is selected so as to avoid this portion of attenuated amplitude. However, JP 8-23359A cannot extract the higher-harmonic component from the flat portion at the crest of the sin/x/waveform but instead extracts it from the sloping portion of the sinx/x waveform. The power of the higher harmonic is therefore low and, moreover, is non-uniform with respect to frequency. This means that the higher-harmonic component cannot be extracted precisely.
The example of the prior art described in the specification of JP 8-149170A is such that since sinx/x-shaped frequency spectrum attenuation is produced in the higher harmonic, a higher harmonic is selected so as to avoid this portion of attenuated amplitude in a manner similar to that of JP 8-23359A.
In other words, JP 8-149170A cannot extract the higher-harmonic component uniformly from the flat portion at the crest of the sinx/x waveform but instead extracts it from the sloping portion of the sinx/x waveform. The power of the higher harmonic is low and non-uniform with respect to frequency and the higher-harmonic component cannot be extracted precisely.
The example of the prior art described in the specification of JP 52-113668A is based upon a frequency conversion operation that utilizes a higher harmonic when an analog signal is processed by a digital filter. However, the output signal of the filter is a sampling pulse that does not undergo zero-order hold. The specification of JP 52-113668A describes sampling pulses using a δ function. In a case where operation is performed with a finite pulse width, a problem similar to that of the prior art shown in FIG. 9 arises.